Method and apparatus for reception in a multi-input-multi-output (MIMO) orthogonal frequency domain modulation (OFDM) wireless communication system

ABSTRACT

An embodiment of the present invention includes a transceiver for use in a multi-input-multi-output (MIMO) Orthogonal Frequency Domain Multiplexing (OFDM) wireless communication system. The transceiver decodes and remodulates certain signal fields and uses the same to update the coefficients of a frequency equalizer thereby improving channel estimation and extending training.

CROSS-REFERENCE TO RELATED APPLICATION

This application is a continuation-in-part of U.S. patent applicationSer. No. 12/632,707, entitled “METHOD AND APPARATUS FOR RECEPTION IN AMULTI-INPUT-MULTI-OUTPUT (MIMO) ORTHOGONAL FREQUENCY DOMAIN MODULATION(OFDM) WIRELESS COMMUNICATION SYSTEM”, filed on Dec. 7, 2009, which is acontinuation-in-part of U.S. patent application Ser. No. 11/475,606 nowU.S. Pat. No. 7,856,068, entitled “NESTED PREAMBLE FOR MULTI INPUT MULTIOUTPUT ORTHOGONAL FREQUENCY DIVISION MULTIPLEXING”, filed on Jun. 26,2006, all of which are incorporated herein by reference.

This application is related to U.S. Pat. No. 7,324,608, entitled“EFFICIENT SUBCARRIER WEIGHTING TO ENHANCE RECEIVER PERFORMANCE”, issuedon Jan. 29, 2008, and U.S. Pat. No. 7,369,626, entitled “EFFICIENTSUBCARRIER EQUALIZATION TO ENHANCE RECEIVER PERFORMANCE”, issued on May6, 2008, all of which are incorporated hereby by reference.

FIELD OF THE INVENTION

The present invention relates generally to the field of wirelesscommunication systems and particularly to a method and apparatus forenhancing channel estimation and subsequent equalization in a receiverused in multi-input-multi-output (MIMO) Orthogonal Frequency DomainMultiplexing Systems (OFDM) wireless communication systems.

BACKGROUND OF THE INVENTION

Communication systems utilize either wire or wireless transmissionaccording to adopted standards. Implementations can range from localwireless networks in the home, to the national and international cellphone networks, to the worldwide Internet.

Communication systems typically conform to one or more of a number ofexisting standards. Wireless standards include the Institute ofElectrical and Electronics Engineers (IEEE) 802.11 wireless local areanetwork (WLAN), the advanced mobile phone services (AMPS), Bluetooth,global system for mobile communications (GSM), code division multipleaccess (CDMA), local multi-point distribution system (LMDS),multi-channel-multi-point distribution systems (MMDS), and variousproprietary implementations of such standards.

Wireless devices in a network, such as a laptop computer, personaldigital assistant, video projector, or WLAN phone, can communicateeither directly or indirectly to other users or devices on the network.In direct communication systems, often referred to as point-to-pointcommunication systems, the two devices are assigned one or morecommunication radio frequency (RF) channels, and the devices communicatedirectly over those channels. In indirect communication systems, thedevices communicate through an intermediary device, such as anassociated base station for cellular services, or an access point forhome or office WLAN networking, on an assigned channel. To complete theconnection, the access point or base station communicates with the pairdirectly, using the system controller, the Public switch telephonenetwork (PSTN), the Internet, or some other wide area network.

It is well known that better channel estimation and equalization canenhance wireless link performance, either through extending the linkrange, or increasing data throughput rates. In noisy channel conditions,performance often suffers because of inaccurate channel estimation, orchannel training. A further problem often encountered is instability inthe analog radio frequency (RF) circuitry, which is often mostpronounced early in the packet, during the channel training portion ofthe preamble. This introduces an error in the channel estimation thatcan lead to packet loss, and degraded link stability. These problems areprevalent in multi-input-multi-output (MIMO) orthogonal frequency domainmodulation (OFDM) communication systems. This is of particular interestin wireless communications systems conforming to the 802.11(n) standardadopted by IEEE because two modes of operation are employed, one beingmixed mode and another being Greenfield.

Methods to improve OFDM channel estimation involve smoothing in thefrequency domain (Perahia 2008), or by using a time-domain formulation(Edfors, et al 2004). The improvement through frequency domain smoothingis limited; particularly in multipath conditions in which case smoothingcan actually degrade performance. The time domain minimum mean-squarederror (MMSE) formulation methods are not desirable because they requireadditional fast fourier transform (FFT)/inverse fast fourier transform(IFFT) modules, which are complex and costly to implement.

Another known method to improve channel estimation/equalization is toimplement an adaptive channel estimation/training algorithm that updatesthe estimate during the data portion of the packet. Adaptive methods canbe complex, and typically take time to converge, during which time anerror can occur.

One simple method to solve the 802.11(n) OFDM channel estimation problemis to use the standard designated training fields in order to performthe channel estimation/equalization function. However, this can besub-optimal, particularly when RF transients corrupt the preambleportion of the packet, or in noisy channel conditions, as much as 2 dBreceiver sensitivity can be lost. Alternately, a frequency domainsmoother can be used to smooth out noise in the channel estimate bytaking advantage of assumed correlation between adjacent subcarriers.However, the smoother can actually degrade performance when thiscorrelation assumption is invalid. Time domain methods are another wayto enhance channel estimation accuracy, but require expensive FFTmodules for each additional receiver chain. Lastly, adaptiveestimation/equalization methods (LMS) can be used to slowly adapt theestimate during the data portion of the packet. However, for shortpackets, the estimate may not converge fast enough to provide anyperformance benefit.

Thus, there exists a need to provide a method and apparatus forimproving channel estimation and subsequent equalization for receiversof multi-input-multi-output (MIMO) orthogonal frequency domainmodulation (OFDM) communication systems that includes a header duringtransmission of information used to decode and enhance the accuracy ofthe channel training.

SUMMARY OF THE INVENTION

Briefly, an embodiment of the present invention includes a transceiverfor use in a multi-input-multi-output (MIMO) Orthogonal Frequency DomainMultiplexing (OFDM) wireless communication system. The transceivertransmits and receives information including signal fields. Thetransceiver includes a first decoder block responsive to a first signalfield of a preamble of a packet transmitted in a communication channel,the first signal field having been modulated prior to being received bythe first decoder. The first decoder block decodes the first signalfield and generates a first decoded preamble field. The transceiverfurther includes a second decoder block of a MIMO OFDM wirelesscommunication system responsive to a second signal field of a preambleof a packet transmitted in a communication channel, the second signalfield having been modulated prior to being received by the seconddecoder. The second decoder block makes a determination as to whetherthe packet is a single stream packet. The second decoder block decodesthe second signal field and generates a second decoded preamble field.The transceiver further includes a first remodulator of a MIMO OFDMwireless communication system responsive to the decoded preamble fieldand operative to generate first remodulated encoded preamble field and afrequency domain equalizer update block operative to update coefficientsof a frequency domain equalizer (FEQ) using the first remodulatedencoded preamble bits. The FEQ update block causes the FEQ to generateequalized preamble field output, which is applied to at least one datafield.

The foregoing and other objects, features and advantages of the presentinvention will be apparent from the following detailed description ofthe preferred embodiments which make reference to several figures of thedrawing.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 shows a multi input multi output (MIMO) transceiver 10, inaccordance with an embodiment of the present invention.

FIG. 2 shows further details of transceiver 10 when used in a mixed modeto send and receive information in accordance with the 802.11(n)standard.

FIG. 3 shows further details of the receiver 28 as well as representingsome of the functions performed by the receiver 28.

FIG. 4 shows a block diagram of the transceiver 10 when used inGreenfield mode, in accordance with an embodiment of the presentinvention.

FIG. 5 shows an alternate embodiment of the present invention where theFEQ, the decoder or demapper are updated.

FIG. 6 shows a graph of the performance of the transceiver 100 in mixedmode and HT with remodulation turned on and off.

FIG. 7 shows a scheme for detecting a large difference between the phasecorrected FFT output from the FFT, between the HT-SIG and HT-LFF fieldsof the packet being received by the transceiver 10, in accordance withanother embodiment of the present invention.

FIG. 8 illustrates a multi-stream packet format showing cyclic shiftdiversity (CSD) on the second stream.

DETAILED DESCRIPTION

Referring now to FIG. 1, a multi input multi output (MIMO) transceiver10 is shown to include a MIMO transmitter 30, an antenna 22, an antenna26, and a receiver 28, in accordance with an embodiment of the presentinvention. The transmitter sends preamble and data information, throughthe antenna 22, to the receiver 28, which receives this informationthrough the antenna 26. A channel model 24, indicated by “h”, representsthe wireless channel between the transmitter 10 and its antenna 22, andthe receiver 28 and its respective antenna 26. Further details ofreceiver 28 are presented and discussed relative to subsequent figures.

Transceiver 10 is shown to include a preamble block 12, an encoder block14, a block interleaver block 16, a mapping block 18 and an inverse fastFourier transform (IFFT) block 20. Preamble block 12 generates preamblebits that are used for equalizer training and synchronization and othertypes of function related to the data that is to be transmittedfollowing the preamble. Included in such information (in bits) isinformation that leads the receiver to detect which mode the transceiver10 is using, i.e. mixed mode or Greenfield, and which type of modulationis used for the data portion of the packet.

Encoder block 14 is shown to receive the preamble bits generated bypreamble block 12 and encodes the same according to known encodingtechniques, an example of which is Viterbi encoding. It is noted thatinformation transmitted by the transmitter 30 is in the form of packetsand the preamble bits are typically organized into fields. Examples offields are provided shortly.

Interleaver block 16 is shown to receive encoded preamble bits fromencoder block 14 and to interleave the same to generate interleavedpreamble bits. The interleaver block 16 utilizes known interleavingtechniques, one such known technique is disclosed in the IEEE802.11(a)-1999 Standard.

Mapping block 18 receives the interleaved preamble bits from interleaverblock 16 and maps the same to orthogonal frequency domain modulation(OFDM) symbols and provides the same to IFFT block 20, which performs aninverse FFT to convert the OFDM symbols to time domain prior totransmission thereof through antenna 22. As previously noted, channelmodel 24 models the channel including the time domain symbols.

Receiver 28 advantageously employs transmitter 30 to remodulate the atleast some of the preamble fields and apply the same to update thecoefficients of its frequency equalizer thereby improving channelestimation and channel training. More specifically, receiver 28advantageously uses the given structure of the OFDM preamble, commonlyknown as “signal fields”, and re-modulates them using transmitter 30,which is an existing and known transmitter, to provide an extendedtraining reference. The advantage is that no additional circuitry isrequired, unlike prior art techniques. Also, it does not require anyassumptions about correlation in the frequency domain betweensubcarrier, so performance improvement with no loss is alwaysguaranteed. Further, the estimate, for example, the channel estimate“h”, is improved in one-shot, at the beginning of the packet, and noconvergence wait time is needed, like the prior art adaptive approach.

It is understood that the figures and discussion presented herein arerelated to a single stream application of transceiver 10 where only onestream is communicated between transmitter 30 and receiver 28. A singlestream can be comprised of 52 subcarriers four of which are pilots(processed by the pilot processor of FIG. 5 in one embodiment of thepresent invention) and the remaining 48 are data for 802.11(a) packets,or 56 subcarriers for 802.11(n) packets, 52 of which are datasubcarriers and 4 reserved for pilots.

FIG. 2 shows further details of transceiver 10 when used in a mixed modeto send and receive information in accordance with the 802.11(n)standard.

Transmitter 30 is shown to perform L-SIG remodulator block 32, HT-SIG0and HT-SIG1 remodulator block 34 and data remodulator block 36, inaccordance with an embodiment of the present invention. Each of theremodulator blocks 32, 34 and 36 includes the blocks shown in thetransmitter 30 of FIG. 1. Alternatively, one or more remodulator may beemployed to carry out the remodulation functions of the individualremodulator blocks 32, 34 and 36. L-SIG, HT-SIG0 and HT-SIG1 areexamples of signal fields. “Remodulator” refers to the functions andstructures used to perform at least encoding, interleaving, mapping.

Receiver 28 is shown to include an FFT block 56, a channelestimation/FEQ block 58, a L-SIG decoder block 60, a HT-SIG decoderblock 62, an update FEQ block 64, a reorder buffer 66, a data decoderblock 68, an data update FEQ block 72 and a data reorder buffer 70.

Channel estimation/FEQ block 58 performs channel estimation and FEQ inthe frequency domain, thus, prior to being processed by the block 58,preamble fields are transformed into the frequency domain, from the timedomain, to ultimately, prior to being applied to one or more datafields, they are transformed from the frequency domain to the timedomain by virtue of an IFFT.

Transmitter 30 sends preamble and data information to receiver 28, aspreviously noted. In FIG. 2, information 38, including preamble and databits, is transmitted by transmitter 30 to receiver 28. In this respect,information 38 is a portion of a mixed mode packet. However, receiver 28cleverly decodes and re-sends certain preamble information back totransmitter 30 for remodulation thereof and use for application to acertain data field(s) thereby extending the training reference andimproving channel estimation. Receiver 28, upon decoding of an HT-SIGfield of information 38 knows that the information it is receiving isfor mixed mode. For further details relating to the two modes, mixed andGreenfield, the reader is referred to in the IEEE P802.11(n)™/D10.00Draft STANDARD for Information Technology-Telecommunications andinformation exchange between systems-Local and metropolitan areanetworks-Specific requirements-Part 11: Wireless LAN Medium AccessControl (MAC) and Physical Layer (PHY) Specifications: Amendment 4:Enhancements for Higher Throughput.

Information 38 is shown to include preamble fields 76 and data fields78. In the mixed mode of 802.11(n), preamble fields 76 include thefollowing fields, which are sent consecutively in the following order:High Throughout (HT)-long training field (HT-LTF0) 40, HT-LTF1 42,legacy (L)-SIG 44, HT-SIG 46, and HT-LTF0 48. Data fields 78 comprisethe following data fields in consecutive order as follows: DATA0 50,DATA1 52 through DATA_N 54.

FFT 56 receives the fields 76 and transforms each field thereof from thetime domain to frequency domain and generates a frequency-domainpreamble for use by the block 58. The block 58 estimates the channel orcalculates an inverse of “h” (the channel model), in frequency domain,and performs frequency equalization on the frequency-domain preamble foruse by the block 60. The block 60 also receive the L-SIG field 44 (“L”representing “long”) and generates decoded preamble bits for use bytransmitter 30, which remodulates the same—L-SIG remodulation isperformed. Similarly, block 62 decodes HT-SIG (“HT” representing HighThroughput) and generates the decoded preamble to transmitter 30—HT-SIGremodulation is performed by transmitter 30. Transmitter 30 generatesremodulated L-SIG and remodulated HT-SIG for use by reorder buffer 66 ofreceiver 28. Thus, the remodulated L-SIG and HT-SIG symbols areadvantageously used as references to update, or improve the initialchannel estimate and FEQ.

Reorder buffer 66 reorders the symbols to reverse the affect of thesymbols having been ordered out of position by the IFFT processperformed by transmitter 30. Reorder buffer 66 generates reorderedremodulated preamble to block 64, which uses the same to update the FEQand to apply the result to DATA1 field 52. While the result of the FEQcan be applied to DATA0 field 50, this is less than desirable because ofthe delay in re-processing the L-SIG field. Further adaptation canfurther improve performance, particularly if the channel parameters arechanging during the coarse of the packet, in which case, this structurecan provide an additional “channel tracking” functionality. It is notedthat the foregoing signal fields (preamble fields) are decoded in timeto apply the same to data thereby improving channel estimation andequalization and decoder weights. Decoder, or demapper weights, fordetermining Viterbi bit metrics, are sometimes referred to as LLR:log-likelihood ratio. Essentially, when the confidence in a particularsubcarrier is small, the corresponding output weights should be small aswell, so that the output from this subcarrier can be adequately“de-weighted” prior to the decoding process. These weights are shown inFIG. 5.

Remodulated L-SIG and HT-SIG preambles are optionally used to update theblock 64 and be applied to DATA2 field 52 through DATA_N fields 54.

In one embodiment of the present invention, the least-means-square (LMS)algorithm, well known to those skilled in the art, is used to update thecoefficients of the FEQ of the block 64. In other embodiments, otherknown techniques, such as but not limited to Recursive Least-squares(RLS), may be employed.

It is noted that HT-SIG field 46 includes HT-SIG0 and HT-SIG1 and bothare provided to the block 62.

In some embodiments, in addition to remodulating the preamble symbols,data symbols are also remodulated and used to update the coefficients ofthe FEQ that equalizes data. For example, in FIG. 2, data fields, suchas DATA_N field 54 is shown provided to the block 68 for decodingthereof. Again, an example of the decoding performed by block 68 isViterbi decoding although decoding is not limited to this type and canbe any suitable type of decoding. The decoded data is then provided byblock 68 to transmitter 30 for remodulation thereof—remodulation 36 isperformed on the data symbols. The remodulated data is then provided bytransmitter 30 to reorder buffer 70, which reorder the data to reversethe disorganizing affects of the transmitter's IFFT and provides thereordered data symbols to block 72, which updates the coefficients ofthe FEQ and equalizes using the remodulated data field (symbol). Thisadditionally improves channel estimation.

It is noted that the HT-LTF0 field 40, HT-LTF1 field 42, L-SIG field 44and HT-SIG field 46 and transmitted as 48 non-pilot (signal) subcarriersand 4 pilot subcarriers and HT-LTF0 field 48 and field 78 aretransmitted as 52 non-pilot subcarriers and 4 pilot subcarriers.

The initial channel estimate 58 (which are used to compute coefficientsof the FEQ) for each of the 52 (including 48 data subcarriers and 4pilot tones) subcarriers, as defined by the 802.11(g) standard, arecalculated in accordance with the following equation:

$\begin{matrix}{{{coef}\; 11g_{k}} = \frac{\sum\limits_{n = 1}^{2}\left( {L - {LTF}_{n}} \right)}{2}} & {{EQ}.\mspace{14mu}(1)}\end{matrix}$

where k is the subcarrier index, and the summation is over the two L-LTFfields. For each kth subcarrier, the entries in the summation are theoutput of the FFT, phase corrected by the training symbol Itf(k) sent onthe particular subcarrier:L−LTF_(n)(k)=fft_out_(n)(k)×ltf(k)  EQ. (2)

The phase correction is simply a pre-stored BPSK symbol, with a value of+1 or −1 depending on the subcarrier number.

The foregoing channel estimation describes a typical prior art receiverchannel estimation 58, employing only the L-LTF portion of the preamble.The prior art FEQ, for each subcarrier, would simply be the inverse ofthe coefficient.

The present invention improves on the prior art receiver by utilizingall additional signal information in the preamble to update the channelcoefficients, and the elements of the FEQ. In one embodiment of theinvention, 802.11(n) implementation, in mixed mode in consideration.Using all signal fields, shown in FIG. 2, the 52 channel coefficientsare calculated in accordance with the following equation:

$\begin{matrix}{{{coef}\; 11g_{k}} = \frac{\begin{matrix}{{HT} - {LTF}_{0} + L - {SIG} +} \\{\sum\limits_{n = 1}^{2}\left( {L - {LTF}_{n} + {HT} - {SIG}_{n}} \right)}\end{matrix}}{6}} & {{EQ}.\mspace{14mu}(3)}\end{matrix}$

where as above L-LTFn are the phase corrected outputs of FFT module 56‘Σ’ represents a summation operation over 2 fields, and the additionalelements HT-SIGn, L-SIG, and HT-LTF0 represent the HT signal field (twototal), the legacy signal field, and the HT long training field,respectively. As for the prior art case, these elements are the FFToutputs of these symbols, and are phase corrected. In order to theproperly phase the L-SIG, HT-SIG FFT outputs, the same symbols need tobe equalized (FEQ 80), decoded (86), and remodulated (32). The output ofthe remodulation block 32 provides BPSK phase for each subcarrier, sothat the L-SIG and HT-SIG fields may be phase corrected and used toimprove the channel estimate. This is a practical solution for manypacket based communication systems containing a preamble/headerstructure because the SIG fields in the header are often modulated atthe lowest modulation rate, making the decoding/remodulation stages fastand accurate.

The mixed-mode 11 n packets are defined with the legacy portionconsisting of 52 total subcarriers. At the start of the HT-STF field(see modified FIG. 2), the signal includes 4 additional subcarriers atthe band edge to increase overall throughput. The additional 4 802.11(n)subcarriers are updated according to the equation:coef11n _(k) =HT−LTF₀  EQ. (4)

This means that the 4 additional subcarriers (two on each band edge)will have less refinement than the 802.11(g) set of 52. Also, Eq. (2)represents the prior art 802.11(n) design, in which only the phasecorrected HT-LTF symbols are used to train all the subcarriers

FIG. 3 shows further details of the receiver 28, and the functionsperformed by the receiver 28 for the refinement of the channelestimation coefficients and FEQ, as described above. In FIG. 3 the block28 performs the typical process of decoding the input signal fields xSIG(x indicating either legacy (L) or HT), shown as inputs to the FEQ 56.The typical receiver block 28 is further shown to include the FEQ block80, a demapper block 82, a de-interleaver block 84, and a decoder block86, in accordance with an embodiment of the present invention. The block58, which computes the initial channel estimation and equalizer, isshown to be coupled to the block 80, which is shown to receive input inthe form of frequency domain (FD) preamble, such as the L-SIG field, andsubsequent data symbols 50 and 52. The block 56 performs a fast Fouriertransform function on the received input and generates afrequency-domain output to the FEQ block 80, which equalizes thefrequency-domain symbols and generates an equalized symbol to thedemapper block 82. The demapper block 82 reverses the affects of theblock 18 done by the transmitter 30 and essentially converts OFDMsymbols to bits for use by the de-interleaver block 84. Thede-interleaver block 84 reverses the affects of the block 16 by thetransmitter 30 and provides de-interleaved preamble to the decoder block86, which decodes the de-interleaved symbols and generates decoded bits(of either the signal or data fields) for use by the transmitter 30 inorder to improve receiver performance. In one embodiment (the receptionof MM packet), the preamble includes the L-SIG field, thus, L-SIGpreamble bits are provided to the block 14 of the transmitter 30 andare, as discussed with reference to FIG. 1, encoded and interleaved andmapped into OFDM symbols with the result being provided to the receiver28. The affect of the foregoing functions performed by the transmitter30 is remodulation of the L-SIG field. The remodulated signal field 81is shown as sig_remo in FIG. 3. The remodulated signal field 81 iscombined with the corresponding stored feq_out 83, from the FFT, to beused by block 63 to update the coefficients of the channel estimation.In this case, the channel update uses these foregoing signals to formthe signal to be used in Eq. (3)L−SIG(k)=fft_out×lsig _(—) remo  EQ. (5)

This process of storing, decoding, and remodulation of the mixed modepreamble is then used for the HT-SIG field to form the HT-SIG element ofEq (3):HT−SIG _(n)(k)=fft_out×htsig _(—) remo  EQ. (6)

To complete the computation of the channel estimate, HT-LTF trainingfield is used. Because this field is a pre-stored training field and notencoded, no remodulation is required to include it into the channelestimation 63. Note that a prior art receiver would only use the HT-LTFfield to estimate the channel.

In effect, the channel estimation of Eq. (3) above is implemented byblock 63 using the remodulated preamble 81 combined with the legacy andHT training fields: L-LTF and HT-LTF. The storage 83, is usedessentially to store the FFT output until the process of remodulatingthe signal field symbols can be remodulated. The remodulated symbols areused to remove the phase of the coded preambles. Finally, using theextended channel estimate, the channel equalizer FEQ can be computed 64.

In FIG. 2, the output of the block 58 is referred to as the frequencydomain (FD) symbol and the remodulation preamble is referred to as theFD reference symbol in FIG. 2.

In some embodiments, remodulation of the signal fields and/or the datafields may be selectively turned ‘on’ and ‘off’ by automaticallydetecting a large difference between the phase corrected FFT outputsfrom the FFT, between the HT-SIG and HT-LTF portions of the packet.Large differences may occur due to effects such as beamforming, orchanges in the cyclic shift delays (as specified in the 802.11(n)standard) that can occur at the start of the HT-STF. One such scheme isdepicted in FIG. 7. In the FIG. 7, the FFT outputs of HT-SIG0 and HT-SIGFields 0 and 1 are phase adjusted using the remodulated correspondingsignal fields, and then subtracted, as shown in FIG. 7. This difference,or error, between the two vectors is formed for each subcarrier,squared, and summed up in the accumulator. This single real number,HT_SIG_diff is stored and used to compare against the same errorcalculation made by subtracting HT-SIG0 and HT-LTF0, HT_LTF_diff. Anyeffect such as beamforming or CSD changes will cause the HT_ LTF diff tobe very much larger (i.e., several times larger) than HT_SIG_diff, whichby definition will not contain any such effects. When the HT_LTF_diff isrelatively large, as shown in FIG. 7, the extended channel estimation,or updating of the channel in block 63 is disabled, and the basicchannel estimation (Eq. 4) is used to perform the estimation andsubsequent equalization. Optionally, in mixed mode, the L_LTF fields areskipped because they can be susceptible to radio frequency (RF)transients that are still from the start of the packet. Thus,optionally, remodulating these fields is not necessary and skipped.

FIG. 4 shows a block diagram of the transceiver 10 when used inGreenfield mode, in accordance with an embodiment of the presentinvention. A transmitted packet will have either a mixed mode structureor a Greenfield structure and the receiver 28 automatically detectswhich type of structure was transmitted and processes the packetsaccordingly. More specifically, this is possible with 802.11(n) devicesbecause there is a 90-degree rotation in the phase of the HT-SIG fieldthat is detectable by the receiver 28, as the preamble fields aremodulated using bipolar phase shift keying (BPSK). In this regard, thepacket is distinguishable. In mixed mode, as in Greenfield mode, thepreamble fields are modulated using BPSK, but the 90-degree shift doesnot occur until after an unrotated L-SIG field, which makes the GF andMM packets uniquely distinguishable.

The structure of the 802.11(n) Greenfield mode packet is similar to theMixed-mode, whereby information 102 is converted into packets, with thedata partitioned into data symbols 78 and the data symbols arepre-pended with preamble fields 110. The fields 110 include a HT-LTF0,HT-LTF1 and HT-SIG fields with the HT-SIG field including an HT-SIG0 andHT-SIG1 fields. Due to the difference in the preambles between theGreenfield and mixed mode and the Greenfield preamble not having a L-SIGfield, only the HT-SIG field is remodulated and used to update the FEQcoefficients when the transceiver 10 is in Greenfield mode. Uponreceiving the remodulated HT-SIG from the transmitter (re-order buffer),the channel coefficients are then updated using three total datasymbols, according to:

$\begin{matrix}{{coef}_{k} = \frac{\sum\limits_{n = 1}^{2}\left( {{HT} - {LTF}_{n} + {HT} - {SIG}_{n}} \right)}{4}} & {{EQ}.\mspace{14mu}(7)}\end{matrix}$

In this equation, the HT-SIG_(n), n=1, 2, are the phase corrected FFToutputs of the HT SIGNAL fields, just as prescribed in Eq (3). Inparticular, the two stored HT-SIGNAL fields received, as stored outputfrom the FFT 83 are phase corrected using the remodulated BPSK symbol81. Similar to the mixed mode case, the HT-LTFn symbols are the FFT 56outputs, phase corrected using the pre-stored BPSK training sequence.

In either mixed mode or Greenfield cases, optionally, data isremodulated and used to update the data FEQ (or the block 72). Using,for example the LMS (least mean squares) algorithm, or simple averagingof the stored, phase corrected FFT outputs. The same hardware (storage,remodulation path circuitry) can be used to implement such an option.This may have advantages because of the additional information that isavailable in the data portion of the packet.

FIG. 5 shows updating of the FEQ block, the decoder and demapper blockweights after updating the channel estimation with the remodulatedsignal field, in an alternative embodiment of the present invention. Thechannel estimation block 58 is shown to provide coefficients to theblock 64 and the pilot processing block 57, which processes the pilotsignals. The block 64 provides updating information to the deriveweights block 59 and the FEQ block 61, which performs the frequencyequalization. The block 59 and the block 61 provide demapper updatinginformation to the demapper block 65, which provides the demapping andprovides the demapped information to the decoder blocks 60, 62 or 68.

Note, in the Figures, blocks 58, 60 and 68 all perform the same decodingfunction. Because these decoding operations can occur at differentnon-overlapping times, it is possible that the same physical decoderperform all the functions depicted by blocks 58, 60 and 68. Re-using thesame decoder block is the preferred embodiment of the present inventionbecause it allows a simpler, less expensive hardware design.

FIG. 6 shows a graph of the performance of the transceiver 100 in mixedmode and HT with remodulation turned on and off. The horizontal orx-axis represents the transmitter's power in a unit of dBm and thevertical or y-axis represents the throughput of the transceiver inmillion bits per second (Mbps). The dashed curve represents theperformance of the transceiver 10 without remodulation and the solidcurve represents the performance of the transceiver 10 withremodulation. This graph shows approximately 4.8 dB improvement in themean square error (MSE) of the channel estimate (H⁻¹), which results inapproximately between 2-3 dB improvement in sensitivity (the horizontalgap between the dashed and solid curves) and this translates to as muchas a 50% increase in throughput (vertical gap distance between thedashed and solid curves).

Extension to Multistream Packets

In particular, reception is improved in a MIMO receiver by fullyutilizing additional information present in the extended preambles thatare part of 802.11n and other forms of wireless communications packets(802.16m, for example). The use of remodulated signal fields (L-SIG, andHT-SIG, where L=legacy and HT=high throughput), and legacy trainingfields (L-LTF) to improve the reception capability of HT packets isdescribed above. The additional information was the remodulated signalfields, combined with the legacy training fields, which were used toimprove the estimate of the channel and subsequent equalization process.The types of packet formats were both Greenfield (GF) and mixed mode(MM), but were limited to single spatial stream packets. Thisdescription covers the extension to multi-stream processing, using thesame approach. As such, the FIGS. 1-7 apply to the extension describedbelow. Finally, the extension to multi-stream processing is importantbecause these formats allow the greatest throughput levels (essentiallydouble) to be achieved by the latest wireless devices, such as 802.11ntransceivers.

The extension to MM 2-stream packets is described hereinbelow, and howthe multi-stream GF is a special case of the MM channel estimateenhancement is also described.

MM Multi-Stream Processing

When two-stream packets are transmitted, a technique known as “cyclicshift diversity” (CSD) will be applied to the second transmitted stream.The cyclic shift, or cyclic delay, is a time shift, either to delay oradvance the second stream with respect to the first. The cyclic shiftprevents unintended interference between the streams. The shifts definedin 802.11n, and shown in the FIG. 8 above, are 200 microseconds, appliedduring legacy portion of the packet (before the HT-STF, short trainingfield), and 400 microseconds is applied to the second stream during theHT portion, after and including the HT-STF.

For MM-HT multi-stream packets, the processing is the same as thatdescribed in FIG. 2: decode the L-SIG (60), remodulate the L-SIG (32),detect the presence of the HT-SIG (46), decode (62) and remodulate theHT-SIG0 and HT-SIG1 (34), and finally, updating the FEQ (64) using theHT-LTF (48), and the remodulated signal fields.

When a multi-stream packet is present, the receiver will become aware ofit during the HT-SIG decoding process (62). When the HT-SIG indicatesthat a two-stream packet is present, previous systems will disregard allthe Legacy fields (signal and training), and use only the receivedHT-LTF fields to estimate the channel, and compute the estimator. Thosefrequency domain signals, Y^(HT0) and Y^(HT1) received during the 1^(st)and 2^(nd) HT-LTF training fields can be expressed as:Y ^(HT0)=(h ₀+δ_(HT) h ₁)X+n ₀Y ^(HT1)=(−h ₀+δ_(HT) h ₁)X+n ₁  EQ. (8)

In this equation, h₀ and h₁ are the channels, X is the sent trainingsymbol, n0, n1 are the measurement noise, and δ_(L) and δ_(HT) are therotations corresponding to the cyclic shifts between the transmitstreams. The latter are complex exponentials written as:δ_(L)=e^(jτ) ^(L) ^(n)δ_(HT)=e^(jτ) ^(HT) ^(n)  EQ. (9)

where τ are the time shift values mentioned above, and n is the OFDMsubcarrier number. Note also that, the first transmitter negates thesecond HT-LTF, which accounts for the negative sign in the Y^(HT1)measurement equation.

The two equations can be expressed in matrix form as:

$\begin{matrix}{Y^{HT} = {{\begin{bmatrix}1 & \delta_{HT} \\{- 1} & \delta_{HT}\end{bmatrix}\begin{bmatrix}h_{0} \\h_{1}\end{bmatrix}}X}} & {{EQ}.\mspace{14mu}(10)}\end{matrix}$

A prior art receiver would lump the cyclic shift parameter into thechannel coefficient vector, simplifying the measurement to:

$\begin{matrix}{Y^{HT} = {{\begin{bmatrix}1 & 1 \\{- 1} & 1\end{bmatrix}\begin{bmatrix}h_{0} \\h_{1}\end{bmatrix}}^{HT}X}} & {{EQ}.\mspace{14mu}(11)}\end{matrix}$

Where, above:

$\begin{matrix}{\begin{bmatrix}h_{0} \\h_{1}\end{bmatrix}^{HT} = \begin{bmatrix}h_{0} \\{\delta_{HT}h_{1}}\end{bmatrix}} & {{EQ}.\mspace{14mu}(12)}\end{matrix}$

A prior art estimate of the channel then is given as:

$\begin{matrix}{\begin{bmatrix}h_{0} \\h_{1}\end{bmatrix}_{HT} = {{\frac{1}{2}\begin{bmatrix}{+ 1} & {- 1} \\{+ 1} & {+ 1}\end{bmatrix}}Y^{HT}}} & {{EQ}.\mspace{14mu}(13)}\end{matrix}$

Note that the prior art estimate does not include any legacy fields. Wecan improve on the estimate in this invention by including the legacyfields, but to do so we must take into account the cyclic shiftdifference between the legacy and HT fields, shown in FIG. 8, throughthe use of a multi-dimensional joint estimate measurement equation.Using the legacy fields, the multi-dimensional joint estimatemeasurement equation can be written as:

$\begin{matrix}{\begin{bmatrix}Y^{L} \\H^{{HT}\; 0} \\Y^{{HT}\; 1}\end{bmatrix} = {{\begin{bmatrix}{+ 1} & {\delta_{L}\delta_{HT}^{*}} \\{+ 1} & 1 \\{- 1} & 1\end{bmatrix}\begin{bmatrix}h_{0} \\h_{1}\end{bmatrix}}_{HT} + N}} & {{EQ}.\mspace{14mu}(14)}\end{matrix}$

Here, the legacy measurements are combined into the vector Y^(L), whichcan be an average of all the previous legacy fields. There are 5 totalfields that can be used, so the average is:

$\begin{matrix}{Y^{L} = {\frac{1}{5}\left( {Y^{L\; 0} + Y^{L\; 1} + Y^{LSIG} + Y^{{HTSIG}\; 0} + Y^{{HTSIG}\; 1}} \right)}} & {{EQ}.\mspace{14mu}(15)}\end{matrix}$

Averaging will improve the estimate because it will reduce the noisevariance on Y^(L) by a factor of square-root of 5 (˜2.31). The improvedHT channel estimate then becomes:

$\begin{matrix}{\begin{bmatrix}h_{0} \\h_{1}\end{bmatrix}_{HT} = {\left( {D^{*}D} \right)^{- 1}{D^{*}\begin{bmatrix}Y^{L} \\Y^{{HT}\; 0} \\Y^{{HT}\; 1}\end{bmatrix}}}} & {{EQ}.\mspace{14mu}(16)}\end{matrix}$

Where

$\begin{matrix}{D = \begin{bmatrix}{+ 1} & {\delta_{L}\delta_{HT}^{*}} \\{+ 1} & 1 \\{- 1} & 1\end{bmatrix}} & {{EQ}.\mspace{14mu}(17)}\end{matrix}$

And

$\begin{matrix}{\left( {D^{*}D} \right)^{- 1} = {\frac{1}{8}\begin{bmatrix}3 & {{- \delta_{L}}\delta_{HT}^{*}} \\{{- \delta_{L}^{*}}\delta_{HT}} & 3\end{bmatrix}}} & {{EQ}.\mspace{14mu}(18)}\end{matrix}$

From these terms, the error covariance of the improved estimate can becalculated, and compared to the prior art estimator. For the prior artestimator, the error variance can be shown to be:

$\begin{matrix}{E_{pa} = {{{trace}\left( {\frac{1}{2}\begin{bmatrix}1 & {- 1} \\1 & 1\end{bmatrix}} \right)} = 1}} & {{EQ}.\mspace{14mu}(19)}\end{matrix}$

Since it is equal to 1, that means that the error variance will be equalto the variance of the received symbols, ie, no estimation gain.

Similarly, the new estimator variance can be shown to be:

$\begin{matrix}\begin{matrix}{E_{new} = {{trace}\left( {D^{*}D} \right)}^{- 1}} \\{= {{trace}\left( {\frac{1}{8}\begin{bmatrix}3 & {{- \delta_{L}}\delta_{HT}^{*}} \\{{- \delta_{L}^{*}}\delta_{HT}} & 3\end{bmatrix}} \right)}} \\{= \frac{3}{4}}\end{matrix} & {{EQ}.\mspace{14mu}(20)}\end{matrix}$

Compared to the conventional estimate, the reduced variance (3/4) willresult in a 1.25 dB improvement in estimate variation. An alternatescheme can further improve on this performance.

Expected MSE Reduction with Redesigned Mixed Mode Estimator

If the entire legacy portion of the preamble is used into one tallvector Y^(L), the row estimate for the mixed mode case has the form:

$\begin{matrix}{{\begin{bmatrix}h_{0} \\h_{1}\end{bmatrix}_{HT} = {\left( {D^{*}D} \right)^{- 1}{D^{*}\begin{bmatrix}\left\lbrack Y^{L} \right\rbrack \\Y^{{HT}\; 0} \\Y^{{HT}\; 1}\end{bmatrix}}}}{D = \begin{bmatrix}{+ 1_{5}} & {\delta_{L}\delta_{HT}^{*}1_{5}} \\{+ 1} & 1 \\{- 1} & 1\end{bmatrix}}} & {{EQ}.\mspace{14mu}(21)}\end{matrix}$

In the estimate, 1₅ is a vector of ones, and the legacy measurementvector is:

$\begin{matrix}{\left\lbrack Y^{L} \right\rbrack = \begin{bmatrix}Y^{L\; 0} \\Y^{L\; 1} \\Y^{LSIG} \\Y^{{HTSIG}\; 0} \\Y^{{HTSIG}\; 1}\end{bmatrix}} & {{EQ}.\mspace{14mu}(22)}\end{matrix}$

In this case, the estimator is:M ⁻¹=(D*D)⁻¹ D*,  EQ. (23)

where

$\begin{matrix}{\left( {D^{*}D} \right)^{- 1} = {{\frac{1}{24}\begin{bmatrix}7 & {{- 5}\;\delta_{L}\delta_{HT}^{*}} \\{{- 5}\;\delta_{L}^{*}\delta_{HT}} & 7\end{bmatrix}}.}} & {{EQ}.\mspace{14mu}(24)}\end{matrix}$

The MSE of the row estimate is then:

$\begin{matrix}\begin{matrix}{{{Tr}\left( {M^{- 1}M^{- *}} \right)} = {{{Tr}\left( {D^{*}D} \right)}^{- 1}{D^{*}\left( {\left( {D^{*}D} \right)^{- 1}D^{*}} \right)}^{*}}} \\{= {{Tr}\left( {D^{*}D} \right)}^{- *}} \\{= {\frac{14}{24}.}}\end{matrix} & {{EQ}.\mspace{14mu}(25)}\end{matrix}$

This provides an MSE reduction of 2.34 dB on each row of the channelmatrix estimate, as compared to the original LTF-only estimate.

Expected MSE Reduction with Redesigned Green Field Mode Estimator

Currently, two HT-LTF0 fields are used, in addition to the HT-LTF1 toestimate the channel. Thus, GF is already better than the LTF only MMestimation. We can quantify this, by writing the estimate as

$\begin{matrix}{{\begin{bmatrix}h_{0} \\h_{1}\end{bmatrix}_{HT} = {\left( {D^{*}D} \right)^{- 1}{D^{*}\begin{bmatrix}Y^{{HT}\; 0} \\Y^{{HT}\; 0} \\Y^{{HT}\; 1}\end{bmatrix}}}}{D = \begin{bmatrix}{+ 1} & {+ 1} \\{+ 1} & {+ 1} \\{- 1} & {+ 1}\end{bmatrix}}} & {{EQ}.\mspace{14mu}(26)}\end{matrix}$

Following the same analysis from above,

$\begin{matrix}{\left( {D^{*}D} \right)^{- 1} = {\frac{1}{8}\begin{bmatrix}3 & {- 1} \\{- 1} & 3\end{bmatrix}}} & {{EQ}.\mspace{14mu}(27)}\end{matrix}$

So, the GF has a MSE on the row estimates of the channel

$\begin{matrix}{{M\; S\; E} = {{{Tr}\left( {D^{*}D} \right)}^{- *} = {\frac{3}{4}.}}} & {{EQ}.\mspace{14mu}(28)}\end{matrix}$

This means that, compared to the MM LTF-only estimate, the GF shouldhave about 1.25 dB better MSE.

GF Estimation Using Remodulated Signal Fields

Again, using the entire legacy portion of the preamble into one tallvector Y^(L), the row estimate for the green field mode case has theform:

$\begin{matrix}{{\begin{bmatrix}h_{0} \\h_{1}\end{bmatrix}_{HT} = {\left( {D^{*}D} \right)^{- 1}{D^{*}\begin{bmatrix}\left\lbrack Y^{L} \right\rbrack \\Y^{{HT}\; 0} \\Y^{{HT}\; 0} \\Y^{{HT}\; 1}\end{bmatrix}}}}{D = \begin{bmatrix}{+ 1_{2}} & {+ 1_{2}} \\{+ 1} & {+ 1} \\{+ 1} & {+ 1} \\{- 1} & {+ 1}\end{bmatrix}}} & {{EQ}.\mspace{14mu}(29)}\end{matrix}$

In the estimate, 1₂ is a vector of ones, and the legacy measurementvector contains only the HTSIG fields:

$\begin{matrix}{\left\lbrack Y^{L} \right\rbrack = \begin{bmatrix}Y^{{HTSIG}\; 0} \\Y^{{HTSIG}\; 1}\end{bmatrix}} & {{EQ}.\mspace{14mu}(30)}\end{matrix}$

Again, the estimator is:M ⁻¹=(D*D)⁻¹ D*,  EQ. (31)

where now:

$\begin{matrix}{\left( {D^{*}D} \right)^{- 1} = {\frac{1}{16}\begin{bmatrix}5 & {- 3} \\{- 3} & 5\end{bmatrix}}} & {{EQ}.\mspace{14mu}(32)}\end{matrix}$

The MSE of the row estimate is then:

$\begin{matrix}{{M\; S\; E} = {{{Tr}\left( {D^{*}D} \right)}^{- *} = {\frac{10}{16}.}}} & {{EQ}.\mspace{14mu}(33)}\end{matrix}$

This results in a MSE reduction of 2.04 dB on the estimate of each rowof the channel matrix.

TABLE 1 Summary of Estimation MSE Improvements (2 Stream Case)Improvement MSE (dB) Mixed Mode No remodulation 1 0 Mixed Mode With14/24 2.34 remodulation Greenfield No remodulation 3/4 1.25 GreenfieldWith 10/16 2.04 remodulationExtension to 3 Stream Processing

The current processing for channel estimation for 3-stream channels, asspecified in the IEEE 802.11n standard, utilizes 4 HT-LTF fields toestimate 3 coefficients per row of the channel matrix. Each receiverchain estimates one row of the matrix.

The row “measurement” can be written as:

$\begin{matrix}{Y^{HT} = {\begin{bmatrix}Y_{0} \\Y_{1} \\Y_{2} \\Y_{3}\end{bmatrix} = {{\begin{bmatrix}{+ 1} & {+ 1} & {+ 1} \\{- 1} & {+ 1} & {+ 1} \\{+ 1} & {- 1} & {+ 1} \\{+ 1} & {+ 1} & {- 1}\end{bmatrix}\begin{bmatrix}h_{0} \\h_{1} \\h_{2}\end{bmatrix}}_{HT} + N}}} & {{EQ}.\mspace{14mu}(34)}\end{matrix}$

If,

$\begin{matrix}{D = \begin{bmatrix}{+ 1} & {+ 1} & {+ 1} \\{- 1} & {+ 1} & {+ 1} \\{+ 1} & {- 1} & {+ 1} \\{+ 1} & {+ 1} & {- 1}\end{bmatrix}} & {{EQ}.\mspace{14mu}(35)}\end{matrix}$

then the row variance estimate can be obtained as above, with

$\begin{matrix}{\left( {D^{*}D} \right)^{- 1} = {\frac{1}{4}\begin{bmatrix}1 & 0 & 0 \\0 & 1 & 0 \\0 & 0 & 1\end{bmatrix}}} & {{EQ}.\mspace{14mu}(36)}\end{matrix}$

Then MSE of the channel row estimate is

$\begin{matrix}{{M\; S\; E} = {{{Tr}\left( {D^{*}D} \right)}^{- *} = {\frac{3}{4}.}}} & {{EQ}.\mspace{14mu}(37)}\end{matrix}$Mixed Mode Case with Extended Estimation Using Remodulated Signal Fields

If we only remodulate the SIGNAL fields and the legacy LTFs, theestimation problem can be expressed as:

$\begin{matrix}{{\begin{bmatrix}h_{0} \\h_{1} \\h_{2}\end{bmatrix}_{HT} = {\left( {D^{*}D} \right)^{- 1}{D^{*}\begin{bmatrix}\left\lbrack Y^{L} \right\rbrack \\Y^{{HT}\; 0} \\Y^{{HT}\; 1} \\Y^{{HT}\; 2} \\Y^{{HT}\; 3}\end{bmatrix}}}}{D = \begin{bmatrix}{+ 1_{5}} & {\delta_{L\; 1}\delta_{H\; 1}^{*}1_{5}} & {\delta_{L\; 2}\delta_{H\; 2}^{*}1_{5}} \\{+ 1} & {+ 1} & {+ 1} \\{- 1} & {+ 1} & {+ 1} \\{+ 1} & {- 1} & {+ 1} \\{+ 1} & {+ 1} & {- 1}\end{bmatrix}}} & {{EQ}.\mspace{14mu}(38)}\end{matrix}$

Here there are 7 elements to average, as opposed to the standard 4HT-LTF's, to estimate the 3 channel coefficients. The inverse portion ofthe estimator has the closed-form solution:

$\begin{matrix}{\left( {D^{*}D} \right)^{- 1} = {\frac{1}{76}\begin{bmatrix}14 & {{- 5}\;\partial_{1}} & {{- 5}\;\partial_{2}} \\{{- 5}\;\partial_{1}^{*}} & 14 & {{- 5}\;{\partial_{1}^{*}\partial_{2}}} \\{{- 5}\;\partial_{1}^{*}} & {{- 5}\;{\partial_{1}\partial_{2}^{*}}} & 14\end{bmatrix}}} & {{EQ}.\mspace{14mu}(39)}\end{matrix}$

Above, the variable ∂₁=δ_(L1)δ*_(H1) is used to simplify notation. So,here the improvement in MSE is given as:

$\begin{matrix}\begin{matrix}{{MSE} = {{Tr}\left( {D^{*}D} \right)}^{- *}} \\{= {\frac{21}{38}.}}\end{matrix} & {{EQ}.\mspace{14mu}(40)}\end{matrix}$

Compared to the standard estimation using only HT-LTFs, the improvementis

$\begin{matrix}{{10\;{\log_{10}\left( \frac{21/38}{3/4} \right)}} = {1.326\mspace{14mu}{{dB}.}}} & {{EQ}.\mspace{14mu}(41)}\end{matrix}$MM Using Only L-SIGNAL and HT-SIGNAL Fields for Remodulation

For a simpler estimate, avoiding L-LTF fields, which may be corrupted byRF transients, or GI errors, the estimation equation can be written

$\begin{matrix}{{\begin{bmatrix}h_{0} \\h_{1} \\h_{2}\end{bmatrix}_{HT} = {\left( {D^{*}D} \right)^{- 1}{D^{*}\begin{bmatrix}\left\lbrack Y^{L} \right\rbrack \\Y^{{HT}\; 0} \\Y^{{HT}\; 1} \\Y^{{HT}\; 2} \\Y^{{HT}\; 3}\end{bmatrix}}}}{D = \begin{bmatrix}{+ 1_{3}} & {\delta_{L\; 1}\delta_{H\; 1}^{*}1_{3}} & {\delta_{L\; 2}\delta_{H\; 2}^{*}1_{3}} \\{+ 1} & {+ 1} & {+ 1} \\{- 1} & {+ 1} & {+ 1} \\{+ 1} & {- 1} & {+ 1} \\{+ 1} & {+ 1} & {- 1}\end{bmatrix}}} & {{EQ}.\mspace{14mu}(42)}\end{matrix}$

Note the portion of the estimate for the legacy shifted signal fieldsnow has dimension 3. Now the inverse portion has the closed form:

$\begin{matrix}{\left( {D^{*}D} \right)^{- 1} = {\frac{1}{52}\begin{bmatrix}10 & {{- 3}\;\partial_{1}} & {{- 3}\;\partial_{2}} \\{{- 3}\;\partial_{1}^{*}} & 10 & {{- 3}\;{\partial_{1}^{*}\partial_{2}}} \\{{- 3}\;\partial_{2}^{*}} & {{- 3}\;{\partial_{1}\partial_{2}^{*}}} & 10\end{bmatrix}}} & {{EQ}.\mspace{14mu}(43)}\end{matrix}$

Again, computing the MSE from the trace function formula

$\begin{matrix}\begin{matrix}{{MSE} = {{Tr}\left( {D^{*}D} \right)}^{- *}} \\{= {\frac{15}{26}.}}\end{matrix} & {{EQ}.\mspace{14mu}(44)}\end{matrix}$

Comparing to the standard, non-remodulated estimate, the improvement forthis mixed-mode case is:

$\begin{matrix}{{10\;{\log_{10}\left( \frac{15/26}{3/4} \right)}} = {1.139\mspace{14mu}{{dB}.}}} & {{EQ}.\mspace{14mu}(45)}\end{matrix}$Greenfield Estimation Using Remodulation, Using Only SIGNAL Fields

For the Greenfield case, the estimation simplifies because there is noCSD transition between preamble and HT data portions of the packet.

$\begin{matrix}{{\begin{bmatrix}h_{0} \\h_{1} \\h_{2}\end{bmatrix}_{HT} = {\left( {D^{*}D} \right)^{- 1}{D^{*}\begin{bmatrix}\left\lbrack Y^{L} \right\rbrack \\Y^{{HT}\; 0} \\Y^{{HT}\; 1} \\Y^{{HT}\; 2} \\Y^{{HT}\; 3}\end{bmatrix}}}}{D = \begin{bmatrix}{+ 1_{2}} & {+ 1_{2}} & {+ 1_{2}} \\{+ 1} & {+ 1} & {+ 1} \\{- 1} & {+ 1} & {+ 1} \\{+ 1} & {- 1} & {+ 1} \\{+ 1} & {+ 1} & {- 1}\end{bmatrix}}} & {{EQ}.\mspace{14mu}(46)}\end{matrix}$

Following the same analysis, the estimator has the closed form solution:

$\begin{matrix}{\left( {D^{*}D} \right)^{- 1} = {\frac{1}{20}\begin{bmatrix}4 & {- 1} & {- 1} \\{- 1} & 4 & {- 1} \\{- 1} & {- 1} & 4\end{bmatrix}}} & {{EQ}.\mspace{14mu}(47)}\end{matrix}$

Then, the MSE is computed as

$\begin{matrix}\begin{matrix}{{MSE} = {{Tr}\left( {D^{*}D} \right)}^{- *}} \\{= {\frac{12}{20}.}}\end{matrix} & {{EQ}.\mspace{14mu}(48)}\end{matrix}$

Similarly, the improvement compared to straight HT-LTF estimation iscomputed as

$\begin{matrix}{{10\;{\log_{10}\left( \frac{21/20}{3/4} \right)}} = {0.97\mspace{14mu}{{dB}.}}} & {{EQ}.\mspace{14mu}(49)}\end{matrix}$

Note, that for GF packets, there are actually 2 HT-LTF0 fields. Weassume that these are averaged together prior to the remodulation andestimation. If we do not pre-average the HT-LTF0 fields, but keep themseparate in the estimate, the D matrix adds a row, by repeating thesecond row, and the form of the estimator is identical to the 7-row MMestimator above, without the phase shift corrections:

$\begin{matrix}{\left( {D^{*}D} \right)^{- 1} = {\frac{1}{52}\begin{bmatrix}10 & {- 3} & {- 3} \\{- 3} & 10 & {- 3} \\{- 3} & {- 3} & 10\end{bmatrix}}} & {{EQ}.\mspace{14mu}(50)}\end{matrix}$

In this case, the MSE improves by 1.139 dB, just as before.

TABLE 2 MSE Improvements for 3-Stream Processing Improvement MSE (dB)Mixed Mode No 3/4 0 remodulation Mixed Mode Remodulation 21/38 1.33 withL-LTF Mixed With 15/26 1.14 remodulation Mode, no L- LTF Greenfield No3/4 0 remodulation Greenfield With 12/20 0.97 remodulation of HT-SIG0,1Summary

Compared to the current MM estimation, the channel estimate can beimproved by up to about 2.3 dB in MSE. The current improvement in GFestimation MSE is up to 0.97 dB.

Note: However, the GF improvement should also provide robustness tocarrier impairments, because the additional HTSIG fields are later inthe preamble portion of the packet.

Although the present invention has been described in accordance with theembodiments shown, one of ordinary skill in the art will readilyrecognize that there could be variations to the embodiments and thosevariations would be within the spirit and scope of the presentinvention. Accordingly, many modifications may be made by one ofordinary skill in the art without departing from the spirit and scope ofthe appended claims.

1. A transceiver for use in a multi-input-multi-output (MIMO) OrthogonalFrequency Domain Multiplexing (OFDM) wireless communication system, thetransceiver operative to transmit and receive information in the form offields, the fields including signal fields, the transceiver comprising:a first decoder block of a MIMO OFDM wireless communication systemresponsive to a first signal field of a preamble of a packet transmittedin a communication channel, the first signal field having been modulatedprior to being received by the first decoder, the first decoder blockconfigured to decode the first signal field and operative to generate afirst decoded preamble field; a second decoder block of a MIMO OFDMwireless communication system responsive to a second signal field of apreamble of a packet transmitted in a communication channel, the secondsignal field having been modulated prior to being received by the seconddecoder, wherein the second decoder block makes a determination as towhether the packet is a single stream packet or a multistream packet;the second decoder block configured to decode the second signal fieldand operative to generate a second decoded preamble field; a firstremodulator of a MIMO OFDM wireless communication system responsive tothe first decoded preamble field and operative to generate firstremodulated encoded preamble field; when the packet is a multistreampacket then the remodulated signal fields are subsequently used toimprove a multidimensional channel estimate; and a frequency equalizerupdate block operative to update coefficients of a frequency equalizer(FEQ) using the first remodulated encoded preamble bits, the FEQ updateblock causing the FEQ to generate equalized data field output, which areapplied to at least one data field, wherein a cyclic shift difference(CSD) between a legacy portion and a high-throughput (HT) portion of themultistream packet is utilized in the estimate.
 2. The transceiver asrecited in claim 1, wherein the second signal field comprises two signalfield symbols.
 3. The transceiver, as recited in claim 2, wherein thetwo signal field symbols are decoded by the second decoder block and areremodulated to provide an improved estimate for use by the FEQ updateblock to further update the coefficients of the FEQ and to apply thesame to the at least one data field.
 4. The transceiver, as recited inclaim 1, wherein the first signal field comprises a legacy signal fieldand the second signal field comprises a high-throughput (HT) signalfield.
 5. The transceiver, as recited in claim 1, wherein the packetcomprises a Greenfield packet.
 6. The transceiver, as recited in claim1, wherein the multi-dimensional joint estimate is provided by theequation: $\begin{bmatrix}h_{0} \\h_{1}\end{bmatrix}_{HT} = {\left( {D^{*}D} \right)^{- 1}{D^{*}\begin{bmatrix}Y^{L} \\Y^{{HT}\; 0} \\Y^{{HT}\; 1}\end{bmatrix}}}$ $D = \begin{bmatrix}{+ 1} & {\delta_{L}\delta_{HT}^{*}} \\{+ 1} & 1 \\{- 1} & 1\end{bmatrix}$ where Y^(L) is the legacy frequency domain signal,Y^(HTO) and Y^(HT1) are high throughput frequency domain signal, h_(o)and h₁ are the channels, δ_(L) and δ_(HT) are the rotationscorresponding to cyclic shifts between the multistreams.
 7. Thetransceiver, as recited in claim 1, wherein the transceiver transmitsand receives information in accordance with the IEEE 802.11(n) standard.8. The transceiver, as recited in claim 1, wherein the coefficients areupdated in accordance with the least means square (LMS) algorithm. 9.The transceiver, as recited in claim 1, wherein the coefficients areupdated in accordance with the Recursive Least-squares (RLS) algorithm.10. The transceiver, as recited in claim 1, wherein the equalizedpreamble field output is applied to more than one data field.
 11. Thetransceiver, as recited in claim 1, further including a data decoderblock of a MIMO OFDM wireless communication system responsive to the atleast one data field and operative to generate decoded data field. 12.The transceiver, as recited in claim 11, further including a dataremodulator of a MIMO OFDM wireless communication system responsive tothe decoded data field and operative to generate remodulated encodeddata field and a data frequency equalizer update block operative toupdate coefficients of a data frequency equalizer (FEQ) using theremodulated encoded data field, the data FEQ update block causing thedata FEQ to generate equalized data field output, which are applied toat least one data field.
 13. The transceiver, as recited in claim 12,wherein said second decoder block is responsive to the second and thirdsignal fields and is operative to generate an error field that iscompared against a threshold and based on the error being larger orsmaller than the threshold, the update block is enabled or disabled. 14.A method of enhancing channel estimation in a multi-input-multi-output(MIMO) Orthogonal Frequency Domain Multiplexing (OFDM) wirelesscommunication system wherein information is transmitted and received inthe form of fields, the fields including signal fields, the methodcomprising: first decoding, in a MIMO OFDM wireless communicationsystem, a first signal field of a preamble of a packet transmitted in acommunication channel, the first signal field having been previouslymodulated, a generating a first decoded preamble field; a seconddecoding, in MIMO OFDM wireless communication system, a second signalfield of a preamble of a packet transmitted in a communication channel,wherein the second decoding makes a determination as to whether thepacket is a single stream packet or a multistream packet the secondsignal field having been previously modulated and generating a seconddecoded preamble field; and a first remodulating, in a MIMO OFDMwireless communication system, when the packet is a multistream packetthen the remodulated signal fields are subsequently used to improve amultidimensional channel estimate; the decoded preamble field and togenerating first remodulated encoded preamble bits; updatingcoefficients of a frequency equalizer (FEQ) using the first remodulatedencoded preamble field and causing the FEQ to generate equalized datafield output, which are applied to at least one data field, wherein acyclic shift difference (CSD) between a legacy portion and ahigh-throughput (HT) portion of the multistream packet is utilized inthe estimate.
 15. The method of claim 14, wherein the first signal fieldcomprises a legacy signal field and the second signal field comprises ahigh-throughput (HT) signal field.
 16. The method of claim 14, whereinthe packet comprises a Greenfield packet.
 17. The method of claim 14,wherein the two-dimensional joint estimate is provided by the equation:$\begin{bmatrix}h_{0} \\h_{1}\end{bmatrix}_{HT} = {\left( {D^{*}D} \right)^{- 1}{D^{*}\begin{bmatrix}Y^{L} \\Y^{{HT}\; 0} \\Y^{{HT}\; 1}\end{bmatrix}}}$ $D = \begin{bmatrix}{+ 1} & {\delta_{L}\delta_{HT}^{*}} \\{+ 1} & 1 \\{- 1} & 1\end{bmatrix}$ where Y^(L) is the legacy frequency domain signal,Y^(HTO) and Y^(HT1) are high throughput frequency domain signal, h_(o)and h₁ are the channels, δ_(L) and δ_(HT) are the rotationscorresponding to cyclic shifts between the multi-streams.
 18. Atransceiver for use in a multi-input-multi-output (MIMO) OrthogonalFrequency Domain Multiplexing (OFDM) wireless communication system, thetransceiver operative to transmit and receive information in the form offields, the fields including signal fields, the transceiver comprising:a first decoder block of a MIMO OFDM wireless communication systemresponsive to a first signal field of a preamble of a packet transmittedin a communication channel, the first signal field having been modulatedprior to being received by the first decoder, the first decoder blockconfigured to decode the first signal field and operative to generate afirst decoded preamble field; a second decoder block of a MIMO OFDMwireless communication system responsive to a second signal field of apreamble of a packet transmitted in a communication channel, the secondsignal field having been modulated prior to being received by the seconddecoder, wherein the second decoder block makes a determination as towhether the packet is a single stream packet or a multistream packet;the second decoder block configured to decode the second signal fieldand operative to generate a second decoded preamble field; a firstremodulator of a MIMO OFDM wireless communication system responsive tothe first decoded preamble field and operative to generate firstremodulated encoded preamble field; when the packet is a multistreampacket then the remodulated signal fields are subsequently used toimprove a multidimensional channel estimate; a frequency equalizerupdate block operative to update coefficients of a frequency equalizer(FEQ) using the first remodulated encoded preamble bits, the FEQ updateblock causing the FEQ to generate equalized data field output, which areapplied to at least one data field, wherein a cyclic shift difference(CSD) between a legacy portion and a high-throughput (HT) portion of themultistream packet is utilized in the estimate; a data decoder block ofa MIMO OFDM wireless communication system responsive to the at least onedata field and operative to generate decoded data field; a dataremodulator of a MIMO OFDM wireless communication system responsive tothe decoded data field and operative to generate remodulated encodeddata field; a data frequency equalizer update block operative to updatecoefficients of a data frequency equalizer (FEQ) using the remodulatedencoded data field, the data FEQ update block causing the data FEQ togenerate equalized data field output, which are applied to at least onedata field, wherein said second decoder block is responsive to thesecond and third signal fields and is operative to generate an errorfield that is compared against a threshold and based on the error beinglarger or smaller than the threshold, the update block is enabled ordisabled.
 19. A method of enhancing channel estimation in amulti-input-multi-output (MIMO) Orthogonal Frequency Domain Multiplexing(OFDM) wireless communication system, wherein information is transmittedand received in the form of fields, the fields including signal fields,the method comprising: first decoding, in a MIMO OFDM wirelesscommunication system, a first signal field of a preamble of a packettransmitted in a communication channel, the first signal field havingbeen previously modulated, a generating a first decoded preamble field;a second decoding, in a MIMO OFDM wireless communication system, asecond signal field of a preamble of a packet transmitted in acommunication channel, wherein the second decoding makes a determinationas to whether the packet is a single stream packet or a multistreampacket the second signal field having been previously modulated andgenerating a second decoded preamble field; and a first remodulating, ina MIMO OFDM wireless communication system, when the packet is amultistream packet then the remodulated signal fields are subsequentlyused to improve a multidimensional channel estimate; the decodedpreamble field and to generating first remodulated encoded preamblebits; updating coefficients of a frequency equalizer (FEQ) using thefirst remodulated encoded preamble field and causing the FEQ to generateequalized data field output, which are applied to at least one datafield, wherein a cyclic shift difference (CSD) between a legacy portionand a high-throughput (HT) portion of the multistream packet is utilizedin the estimate, data decoding in a MIMO OFDM wireless communicationsystem responsive to the at least one data field and operative togenerate decoded data field, a data remodulating in a MIMO OFDM wirelesscommunication system responsive to the decoded data field and operativeto generate remodulated encoded data field; and a data frequencyequalizer updating operative to update coefficients of a data frequencyequalizer (FEQ) using the remodulated encoded data field, the data FEQupdating causing the data FEQ to generate equalized data field output,which are applied to at least one data field, wherein said seconddecoding is responsive to the second and third signal fields and isoperative to generate an error field that is compared against athreshold and based on the error being larger or smaller than thethreshold, the updating is enabled or disabled.
 20. A transceiver foruse in a multi-input-multi-output (MIMO) Orthogonal Frequency DomainMultiplexing (OFDM) wireless communication system, the transceiveroperative to transmit and receive information in the form of fields, thefields including signal fields, the transceiver comprising: a firstdecoder block of a MIMO OFDM wireless communication system responsive toa first signal field of a preamble of a packet transmitted in acommunication channel, the first signal field having been modulatedprior to being received by the first decoder, the first decoder blockconfigured to decode the first signal field and operative to generate afirst decoded preamble field; a second decoder block of a MIMO OFDMwireless communication system responsive to a second signal field of apreamble of a packet transmitted in a communication channel, the secondsignal field having been modulated prior to being received by the seconddecoder, wherein the second decoder block makes a determination as towhether the packet is a single stream packet or a multistream packet;the second decoder block configured to decode the second signal fieldand operative to generate a second decoded preamble field; a firstremodulator of a MIMO OFDM wireless communication system responsive tothe first decoded preamble field and operative to generate firstremodulated encoded preamble field; when the packet is a multistreampacket then the remodulated signal fields are subsequently used toimprove a multidimensional channel estimate; and a frequency equalizerupdate block operative to update coefficients of a frequency equalizer(FEQ) using the first remodulated encoded preamble bits, the FEQequalizer update block causing the FEQ to generate equalized data fieldoutput, which are applied to at least one data field, wherein themulti-dimensional channel estimate is provided by the equation:$\begin{bmatrix}h_{0} \\h_{1}\end{bmatrix}_{HT} = {{\left( {D^{*}D} \right)^{- 1}{D^{*}\begin{bmatrix}Y^{L} \\Y^{{HT}\; 0} \\Y^{{HT}\; 1}\end{bmatrix}}\mspace{14mu} D} = \begin{bmatrix}{+ 1} & {\delta_{L}\delta_{HT}^{*}} \\{+ 1} & 1 \\{- 1} & 1\end{bmatrix}}$ where Y^(L) is the legacy frequency domain signal,Y^(HTO) and Y^(HT1) are high throughput frequency domain signal, h_(o)and h₁ are the channels, δ_(L) and δ_(HT) are the rotationscorresponding to cyclic shifts between the multistreams.
 21. A method ofenhancing channel estimation in a multi-input-multi-output (MIMO)Orthogonal Frequency Domain Multiplexing (OFDM) wireless communicationsystem wherein information is transmitted and received in the form offields, the fields including signal fields, the method comprising: firstdecoding, in a MIMO OFDM wireless communication system, a first signalfield of a preamble of a packet transmitted in a communication channel,the first signal field having been previously modulated, a generating afirst decoded preamble field; a second decoding, in MIMO OFDM wirelesscommunication system, a second signal field of a preamble of a packettransmitted in a communication channel, wherein the second decodingmakes a determination as to whether the packet is a single stream packetor a multistream packet the second signal field having been previouslymodulated and generating a second decoded preamble field; and a firstremodulating, in a MIMO OFDM wireless communication system, when thepacket is a multistream packet then the remodulated signal fields aresubsequently used to improve a multidimensional channel estimate; thedecoded preamble field and to generating first remodulated encodedpreamble bits; updating coefficients of a frequency equalizer (FEQ)using the first remodulated encoded preamble field and causing the FEQto generate equalized data field output, which are applied to at leastone data field, wherein a cyclic shift difference (CSD) between a legacyportion and a high-throughput (HT) portion of the multistream packet isutilized in the estimate, wherein the two-dimensional channel estimateis provided by the equation: $D = {{\begin{bmatrix}{+ 1} & {\delta_{L}\delta_{HT}^{*}} \\{+ 1} & 1 \\{- 1} & 1\end{bmatrix}\mspace{14mu}\begin{bmatrix}h_{0} \\h_{1}\end{bmatrix}}_{HT} = {\left( {D^{*}D} \right)^{- 1}{D^{*}\begin{bmatrix}Y^{L} \\Y^{{HT}\; 0} \\Y^{{HT}\; 1}\end{bmatrix}}}}$ where Y^(L) is the legacy frequency domain signal,Y^(HTO) and Y^(HT1) are high throughput frequency domain signal, h_(o)and h₁ are the channels, δ_(L) and δ_(HT) are the rotationscorresponding to cyclic shifts between the multi-streams.